The Litvak Legacy

William Allan Kritsonis, PhD

 

Dr. Kritsonis Lectures at the University of Oxford, Oxford, England

 

In 2005, Dr. Kritsonis was an Invited Visiting Lecturer at the Oxford Round Table at Oriel College in the University of Oxford, Oxford, England.  His lecture was entitled the Ways of Knowing Through the Realms of Meaning.

 

Dr. Kritsonis Recognized as Distinguished Alumnus

 

         In 2004, Dr. William Allan Kritsonis was recognized as the Central Washington University Alumni Association Distinguished Alumnus for the College of Education and Professional Studies.  Dr. Kritsonis was nominated by alumni, former students, friends, faculty, and staff.  Final selection was made by the Alumni Association Board of Directors.  Recipients are CWU graduates of 20 years or more and are recognized for achievement in their professional field and have made a positive contribution to society.  For the second consecutive year, U.S. News and World Report placed Central Washington University among the top elite public institutions in the west.  CWU was 12th on the list in the 2006 On-Line Education of “America’s Best Colleges.”

  

 

Educational Background

 

          Dr. William Allan Kritsonis earned his BA in 1969 from Central Washington University, Ellensburg, Washington.  In 1971, he earned his M.Ed. from Seattle Pacific University.  In 1976, he earned his PhD from the University of Iowa.  In 1981, he was a Visiting Scholar at Teachers College, Columbia University, New York, and in 1987 was a Visiting Scholar at Stanford University, Palo Alto, California. In June 2008, Dr. Kritsonis received the Doctor of Humane Letters, School of Graduate Studies from Southern Christian University. The ceremony was held at the Hilton Hotel in New Orleans, Louisiana.

 

Professional Experience

 

Dr. Kritsonis began his career as a teacher.  He has served education as a principal, superintendent of schools, director of student teaching and field experiences, invited guest professor, author, consultant, editor-in-chief, and publisher.  Dr. Kritsonis has earned tenure as a professor at the highest academic rank at two major universities.

 

Books – Articles – Lectures – Workshops

 

     Dr. Kritsonis lectures and conducts seminars and workshops on a variety of topics.  He is author of more than 500 articles in professional journals and several books.  His popular book SCHOOL DISCIPLINE: The Art of Survival is scheduled for its fourth edition.  He is the author of the textbook William Kritsonis, PhD on Schooling that is used by many professors at colleges and universities throughout the nation and abroad. 

 In 2007, Dr. Kritsonis’ version of the book of Ways of Knowing Through the Realms of Meaning (858 pages) was published in the United States of America in cooperation with partial financial support of Visiting Lecturers, Oxford Round Table (2005).  The book is the product of a collaborative twenty-four year effort started in 1978 with the late Dr. Philip H. Phenix.  Dr. Kritsonis was in continuous communication with Dr. Phenix until his death in 2002.

 In 2007, Dr. Kritsonis was the lead author of the textbook Practical Applications of Educational Research and Basic Statistics.  The text provides practical content knowledge in research for graduate students at the doctoral and master’s levels.

            In 2008, Dr. Kritsonis’ book Non-Renewal of Public School Personnel Contracts: Selected Supreme and District Court Decisions in Accordance with the Due Process of Law is scheduled for publication by The Edwin Mellen Press, Lewiston, New York.

           Dr. Kritsonis’ seminar and workshop on Writing for Professional Publication has been very popular with both professors and practitioners.  Persons in attendance generate an article to be published in a refereed journal at the national or international levels.

           Dr. Kritsonis has traveled and lectured throughout the United States and world-wide.  Some recent international tours include Australia, New Zealand, Tasmania, Turkey, Italy, Greece, Monte Carlo, England, Holland, Denmark, Sweden, Finland, Russia, Estonia, Poland, Germany, and many more.

 

 

 Founder of National FORUM Journals – Over 4,000 Professors Published

 

          Dr. Kritsonis is founder of NATIONAL FORUM JOURNALS (since 1983).  These publications represent a group of highly respected scholarly academic periodicals.  Over 4,000 writers have been published in these refereed, peer-reviewed periodicals.  In 1983, he founded the National FORUM of Educational Administration and Supervision – now acclaimed by many as the United States’ leading recognized scholarly academic refereed journal in educational administration, leadership, and supervision.

          In 1987, Dr. Kritsonis founded the National FORUM of Applied Educational Research Journal whose aim is to conjoin the efforts of applied educational researchers world-wide with those of practitioners in education.  He founded the National FORUM of Teacher Education Journal, National FORUM of Special Education Journal, National FORUM of Multicultural Issues Journal, International Journal of Scholarly Academic Intellectual Diversity, International Journal of Management, Business, and Administration, and the DOCTORAL FORUM – National Journal for Publishing and Mentoring Doctoral Student Research. The DOCTORAL FORUM is the only refereed journal in America committed to publishing doctoral students while they are enrolled in course work in their doctoral programs. In 1997, he established the Online Journal Division of National FORUM Journals that publishes academic scholarly refereed articles daily on the website: www.nationalforum.com.  Over 500 professors have published online.  In January 2007, Dr. Kritsonis established Focus: On Colleges, Universities, and Schools.

 

Professorial Roles

 

          Dr. Kritsonis has served in professorial roles at Central Washington University, Washington; Salisbury State University, Maryland; Northwestern State University, Louisiana; McNeese State University, Louisiana; and Louisiana State University, Baton Rouge in the Department of Administrative and Foundational Services.                                          

     In 2006, Dr. Kritsonis published two articles in the Two-Volume Set of the Encyclopedia of Educational Leadership and Administration published by SAGE Publications, Thousand Oaks, California. He is a National Reviewer for the Journal of Research on Leadership, University Council for Educational Administration (UCEA).

     In 2007, Dr. Kritsonis has been invited to write a history and philosophy of education for the ABC-CLIO Encyclopedia of World History.

           Currently, Dr. Kritsonis is Professor of Educational Leadership at Prairie Vi
ew A&M University – Member of the Texas A&M University System.  He teaches in the newly established PhD Program in Educational Leadership.  Dr. Kritsonis taught the Inaugural class session in the doctoral program at the start of the fall 2004 academic year. In October 2006, Dr. Kritsonis chaired the first doctoral student to earn a PhD in Educational Leadership at Prairie View A&M University.  He lives in Houston, Texas.

Instructions for Oral Care during chemotherapy

It is necessary that you understand the importance of good oral hygiene and preventive care before, during, and after chemotherapy.

The purpose of chemotherapy is to reduce the growth of tumor cells.  Unfortunately, tissues in the mouth are affected by chemotherapy and special care should be taken to help prevent infections in the mouth.

Ways of preventing these problems start with an examination by a dentist.  If any dental procedures are necessary before chemotherapy, they usually can be done while the bone can heal properly.

For more information and photos check our site: www.penchasdentistry.com

Common Side Effects

Dry Mouth is very common after the first week of therapy and may persist as a permanent effect.  This depends on how and what type of drugs are given.  This will vary from patient to patient. Mouth Ulcers are a common occurrence during chemotherapy.  Sometimes, these ulcers are preventable by simple oral hygiene care and a cautious diet.  These may occur on the lips, tongue, and roof of mouth and inside the check. Swollen gums are another side effect that may occur if tartar has accumulated around and below the gum line of your teeth. Dental Abscesses can occur if broken or impacted teeth are present during chemotherapy.

Oral Hygiene

A soft toothbrush held like a pen can be used carefully during chemotherapy.  If the mouth is too sore to use a toothbrush, a folded square gauze pad can be moistened with water and gently wiped over the surfaces of the teeth and/ or gums instead.

Rinsing the mouth with a salt and soda solution will remove food and debris which tends to accumulate in the mouth.  (This is made by adding one teaspoon of salt and one teaspoon of baking soda to one quart of water).  This solution can be used as often as necessary to provide a clean oral environment.

Things to avoid

Avoid foods that have a rough consistency such as:  tortilla chips, fried foods, nuts, etc. Avoid wearing complete dentures during chemotherapy, as this will promote nausea and vomiting.  Wearing dentures will also promote development of mouth ulcers Avoid a prolonged dry mouth which may tend to develop during sleep.  Keep plenty of water at bedside. Avoid carbonated beverages and juices that are acidic like grapefruit or orange juice. Avoid high alcohol containing mouthwashes like Listerine or Scope.  Use only the salt and soda solution. If vomiting occurs, rinse with soda and salt solution; this will neutralize the stomach acid.  Prolonged contact of stomach acid in the mouth will cause mouth ulcers.

A novel analysis of energy efficiency motors and power controllers

Voltage Control

Voltage alone can be used as a source of intelligence when the switched capacitors are applied at point where the circuit voltage decreases as circuit load increases. Generally, where they are applied the voltage should decrease as circuit load increases and the drop in voltage should be around 4 – 5 % with increasing load.

Voltage is the most common type of intelligence used in substation applications, when maintaining a particular voltage is of prime importance. This type of control is independent of load cycle. During light load time and low source voltage, this may give leading PF at the substation, which is to be taken note of. KILOVAR Control

Automatic Power Factor Control Relay

It controls the power factor of the installation by giving signals to switch on or off power factor correction capacitors. Relay is the brain of control circuit and needs contactors of appropriate rating for switching on/off the capacitors.

There is a built-in power factor transducer, which measures the power factor of the installation and converts it to a DC voltage of appropriate polarity. This is compared with a reference voltage, which can be set by means of a knob calibrated in terms of power factor.

When the power factor falls below setting, the capacitors are switched on in sequence. The relays are provided with First in First out (FIFO) and First in Last Out (FILO) sequence. The capacitors controlled by the relay must be of the same rating and they are switched on/off in linear sequence. To prevent over correction hunting, a dead band is provided. This setting determines the range of phase angle over which the relay does not respond; only when the PF goes beyond this range, the relay acts. When the load is low, the effect of the capacitors is more pronounced and may lead to hunting. Under current blocking (low current cut out) shuts off the relay, switching off all capacitors one by one in sequence, when load current is below setting. Special timing sequences ensure that capacitors are fully discharged before they are switched in. This avoids dangerous over voltage transient. The solid state indicating lamps (LEDS) display various functions that the operator should know and also and indicate each capacitor switching stage.

Intelligent Power Factor Controller (IPFC)

This controller determines the rating of capacitance connected in each step during the first hour of its operation and stores them in memory. Based on this measurement, the IPFC switches on the most appropriate steps, thus eliminating the hunting problems normally associated with capacitor switching.

Energy Efficient Motors

Minimising Watts Loss in Motors

Improvements in motor efficiency can be achieved without compromising motor performance – at higher cost – within the limits of existing design and manufacturing technology.

From the Table .1, it can be seen that any improvement in motor efficiency must result from reducing the Watts losses. In terms of the existing state of electric motor technology, a reduction in watts losses can be achieved in various ways.

All of these changes to reduce motor losses are possible with existing motor design and manufacturing technology. They would, however, require additional materials and/or the use of higher quality materials and improved manufacturing processes resulting in increased motor cost.

Energy Efficient Motor

Table 1

Thus energy-efficient electric motors reduce energy losses through improved design, better materials, and improved manufacturing techniques. Replacing a motor may be justifiable solely on the electricity cost savings derived from an energy-efficient replacement. This is true if the motor runs continuously, power rates are high, the motor is oversized for the application, or its nominal efficiency has been reduced by damage or previous rewinds. Efficiency comparison for standard and high efficiency motors is shown in Figure 2.

Fig.2

Technical aspect of energy efficiency motors

Energy-efficient motors last longer, and may require less maintenance. At lower temperatures, bearing grease lasts longer; required time between re-greasing increases. Lower temperatures translate to long lasting insulation. Generally, motor life doubles for each 10°C reduction in operating temperature.

Select energy-efficient motors with a 1.15 service factor, and design for operation at 85% of the rated motor load.

Electrical power problems, especially poor incoming power quality can affect the operation of energy-efficient motors.

Speed control is crucial in some applications. In polyphase induction motors, slip is a measure of motor winding losses. The lower the slip, the higher the efficiency. Less slippage in energy efficient motors results in speeds about 1% faster than in standard counterparts.

Starting torque for efficient motors may be lower than for standard motors. Facility managers should be careful when applying efficient motors to high torque applications.

Soft Starter

When starting, AC Induction motor develops more torque than is required at full speed. This stress is transferred to the mechanical transmission system resulting in excessive wear and premature failure of chains, belts, gears, mechanical seals, etc. Additionally, rapid acceleration also has a massive impact on electricity supply charges with high inrush currents drawing +600% of the normal run current.

Soft Starter

The use of Star Delta only provides a partial solution to the problem. Should the motor slow down during the transition period, the high peaks can be repeated and can even exceed direct on line current. Soft starter (see Figure 10.5) provides a reliable and economical solution to these problems by delivering a controlled release of power to the motor, thereby providing smooth, stepless acceleration and deceleration. Motor life will be extended as damage to windings and bearings is reduced. Soft Start & Soft Stop is built into 3 phase units, providing controlled starting and stopping with a selection of ramp times and current limit settings to suit all applications

Soft Starter: Starting current, Stress profile during starting

Advantages of Soft Start –

Less mechanical stress

Improved power factor

Lower maximum demand

Less mechanical maintenance

There is a variety of Scholarships Women Minorities you can apply for and you can easily find them online.  We searched for Scholarships Women Minorities and here’s a few we came across that you need to apply for.  Remember to do your own research online for even more scholarship opportunities.

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One of these scholarship opportunities is the $10K scholarship giveaway by freecollegescholarships.net.  To apply for this scholarship you only need to be eighteen years old and be a resident of the United States.  A quick and easy registration form on their site is all it takes for you to enter the scholarship drawing.  There is a deadline for the scholarship drawing so remember to register as soon as possible.

You can learn more about the Higher Reach Scholarship program.  Walmart associates have a chance to get $13,000 of money over 4 years.  The only term is that those students who receive the fellowship need to work at the writing center for three of their four academic years.

You may also be able to qualify for a scholarship program called the Carnegie Writing Scholarship.  Any program of study can be funded with the $1,500 from the scholarship program.  Two requirements to apply are that you need to show financial need and have at least a 2.5 grade point average.

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To make the best of your scholarship application search process, make sure to apply to as many scholarship programs as you can.  You control the success you will have in college.

Remember to apply to as many scholarship programs as you can but only for those you qualify for so you will not be wasting your time.  The more of them that you apply for, the more chances you will have of winning scholarships.

Uniqueness in Common

A common characteristic among fashion designers and those employed within the fashion industry is their uniqueness. That sounds strange, doesn’t it? But think about it: those in the fashion industry are unique individuals, and the top fashion design colleges recognize this quality and offer educational and training programs to help develop the skills of their fashion design students. Whether you want to study fashion design from a location close to your home or in a large fashion design metropolis, like New York or San Francisco, top fashion design colleges that will meet your educational needs are located throughout the United States.

You Have to Start Somewhere

Although most fashion designers have a passion for style and design, few are born with an innate ability to design clothes and apparel. Even the world’s leading designers had to learn and develop their skills before realizing success within the fashion industry. They sought training and education from the top fashion design colleges to develop their designing skills and express their visions. Choosing a fashion school, one that fits your particular uniqueness, is very important to experiencing longer-term success in the industry of fashion.

Fashion designers are artistic people who hold within a desire to create new and exciting designs. Combining knowledge and artistic ability, fashion designers turn their ideas into real fashions for merchandising, publications, and the interiors of the homes we inhabit. Many fashion designers are self-employed and provide fashion services for individual clients, while others provide similar services to department and specialty stores. Designers working for apparel manufacturers generally modify fashions created by other designers to meet the needs of mass marketing.

Keep Up With the Changes

Fashion design is a constantly changing industry, and therefore so are its occupations. The curricula of the top fashion design colleges are structured to provide a well-rounded and contemporary educational experience to the fashion design student. Students can learn the most current innovations taking place in fashion design. They are exposed to computer generated fashion design, pattern development and drafting, fashion merchandising, and much more. If you are interested in a career of fashion design, then you should investigate the courses of study from some of the top fashion design colleges in the nation.

Top Fashion Colleges and World Travel

Most top fashion design colleges provide programs that combine traditional skills – like sewing, pattern production and use of fabric – with the standards and trends of today’s fashions. In any one of these colleges, the work performed is geared toward assisting students to build their own fashion design portfolio. Some top fashion design colleges even provide study tours of the world’s leading fashion producers located in cities like New York and Paris. Imagine traveling to Paris, as a student, to study the styles and techniques of some of the world’s most respected designers. You might even have an opportunity to experience the thrill of one of your own designs being displayed on the runway. It’s all possible if you graduate from one of the top fashion colleges!

To become great human being there is requirement of education. The best educations are given to students in universities that are so famous and worldwide recognized. UAE is developed emirates and have lots of universities and colleges there in major cities like Dubai, Abu Dhabi. Universities of UAE are well known recognized by world wide. These universities give good facilities to their students. And have positive response in result and appreciated by worldwide. There are some universities in UAE along with colleges, described below:

Abu Dhabi University: This is founded at date of 13 of September and owned by University of Abu Dhabi Board of Trustees and Governors. This is private university and located in Khafila city with PO Box 59911, Abu Dhabi, UAE along with telephone number +971-2-5015555 and fax +971-2-5015990. The website for university is www.adu.ac.ae.

Ajman University of Science and Technology: This is founded at 1988 year and directed by Dr Saeed Abdalla Salman (President). Approximate students number are 11,000 with type of university and Ministry of Higher Education license. The location for this is university is PO Box 346, Ajman, UAE along with phone number +971-6-7466666 and fax +971-6-7482277. Website for this university is www.ajman.ac.ae .

Al Ghurair University: This is founded at 1999 and directed by Abdurahim Mohammed Al Ameen (President). This is private university and Ministry of Higher Education licensed. Location of this university is PO Box 37374, Dubai, UAE along with +971-4-4200223 and fax +971-4-4200224. website for this university is www.agu.ae .

Al Hosn University: This is founded at September 2005 and directed by Professor Abdul Rahim Sabouni and owned by Abu Dhabi Holding Company / Board of Trustees. Number of students approximate is 500 and this is private type of university. Location for this university is PO Box 38772, Abu Dhabi, UAE along with telephone number +971-2-4070700 and fax number +971-2-4070799. Website for this university is www.alhosnu.ae

Dubai Aerospace University: This is founded at 2006, directed by Dr George H Ebbs (CEO) and owned by Dubai Aerospace Enterprise DAE. This is one of private company. Location of this university is PO Box 506591, Dubai, UAE along with fax number +971-4-4038189.

French Fashion University: This is founded at 2006 and directed by Denis Ravizza (Director). This is one of private universities. Location of university is Block 4, Dubai Academic City, PO Box 211021, Dubai, UAE along with phone number +971-4-4291228 and fax number +971-4-3604833. Website for this university is www.french-fashion-university.com .

Hamdan eTQM University: This is founded at September 2002, directed by Dr Mansoor Al Awar (Vice-President & CEO) and owned by Board of Governors. This is one of Ministry of Higher Education licensed universities. Location of this university is 1st Floor, Bur Dubai Traffic Department, Sheikh Zayed Road, Al Barsha, PO Box 71400, Dubai, UAE along with +971-4-4088405 and fax number +971-4-4088555. Website for university is www.hbmeu.ae.

Murdoch University Dubai: This is founded at September 2008, directed by Professor J Michael Innes (Dean / Pro-Vice Chancellor). This is one of private university. Location of this university is Block 10, 4th floor, Dubai International Academic City, PO Box 502971, Dubai, UAE along with +971-4-4355700 and fax number +971-4-4264708. Website for this university is www.murdochdubai.com .

visit website for dubai universities , uae universities

There is good reason to run a stepping motor at a supply voltage above that needed to push the maximum rated current through the motor windings. Running a motor at higher voltages leads to a faster rise in the current through the windings when they are turned on, and this, in turn, leads to a higher cutoff speed for the motor and higher torques at speeds above the cutoff.

 

Microstepping, where the control system positions the motor rotor between half steps, also requires external current limiting circuitry. For example, to position the rotor 1/4 of the way from one step to another, it might be necessary to run one motor winding at full current while the other is run at approximately 1/3 of that current.

 

The remainder of this section discusses various circuits for limiting the current through the windings of a stepping motor, starting with simple resistive limiters and moving up to choppers and other switching regulators. Most of these current limiters are appropriate for many other applications, including limiting the current through conventional DC motors and other inductive loads.

 

 

Resistive Current Limiters

The easiest to understand current limiter is a series resistor. Most motor manufacturers recommended this approach in their literature up until the early 1980′s, and most motor data sheets still give performance curves for motors driven by such circuits. The typical circuits used to control the current through one winding of a permanent magnet or hybrid motor are shown in Figure 4.1.

 

Figure 4.1

R1 in this figure limits the current through the motor winding. Given a rated current of I and a motor winding with a resistance Rw, Ohm’s law sets the maximum supply voltage as I(Rw+R1). Given that the inductance of the motor motor winding is Lw, the time constant for the motor winding will be Lw/(Rw+R1). Figure 4.2 illustrates the effect of increasing the resistance and the operating voltage on the rise and fall times of the current through one winding of a stepping motor.

Figure 4.2

R2 is shown only in the unipolar example in Figure 4.1 because it is particularly useful there. For a bipolar H-bridge drive, when all switches are turned off, current flows from ground to the motor supply through R1, so the current through the motor winding will decay quite quickly. In the unipolar case, R2 is necessary to equal this performance. When the switches in the H-bridge circuit shown in Figure 4.1 are opened, the direction of current flow through R1 will reverse almost instantaneously! If R1 has any inductance, for example, if it is wire-wound, it must either be bypassed with a capacitor to handle the voltage kick caused by this current reversal, or R2 must be added to the H-bridge.

 

Given the rated maximum current through each winding and the supply voltage, the resistance and wattage of R1 is easy to compute. R2 if it is included, poses more interesting problems. The resistance of R2 depends on the maximum voltage the switches can handle. For example, if the supply voltage is 24 volts, and the switches are rated at 75 volts, the drop across R2 can be as much as 51 volts without harming the transistors. Given an operating current of 1.5 amps, R2 can be a 34 ohm resistor. Note that an interesting alternative is to use a zener diode in place of R2.

 

Figuring the peak average power R2 must dissipate is a wonderful exercise in dynamics; the inductance of the motor windings is frequently undocumented and may vary with the rotor position. The power dissipated in R2 also depends on the control system. The worst case occurs when the control system chops the power to one winding at a high enough frequency that the current through the motor winding is effectively constant; the maximum power is then a function of the duty cycle of the chopper and the ratios of the resistances in the circuit during the on and off phases of the chopper. Under normal operating conditions, the peak power dissipation will be significantly lower.

 

 

Linear Current Limiters

 

A pair of high wattage power resistors can cost more than a pair of power transistors plus a heat sink, particularly if forced air cooling is available. Furthermore, a transistorized constant current source, as shown in Figure 4.3, will give faster rise times through the motor windings than the current limiting resistor shown in Figure 4.1. This is because a current source will deliver the full supply voltage across the motor winding until the current reaches the rated current; only then will the current source drop the voltage.

 

 

 

Figure 4.3

In Figure 4.3, a transistorized current source (T1 plus R1) has been substituted for the current limiting resistor R1 used in the examples in Figure 4.1. The regulated voltage supplied to the base of T1 serves to regulate the voltage across the sense resistor R1, and this, in turn, maintains a constant current through R1 so long as any current is allowed to flow through the motor winding. Typically, R1 will have as low a resistance as possible, in order to avoid the high cost of a power resistor. For example, if the forward voltage drops across the diode in series with the base T1 and VBE for T1 are both 0.65 volts, and if a 3.3 volt zener diode is used for a reference, the voltage across R1 will be maintained at about 2.0 volts, so if R1 is 2 ohms, this circuit will limit the current to 1 amp, and R1 must be able to handle 2 watts. R3 in Figure 4.3 must be sized in terms of the current gain of T1 so that sufficient current flows through R1 and R3 to allow T1 to conduct the full rated motor current.

 

The transistor T1 used as a current regulator in Figure 4.3 is run in linear mode, and therfore, it must dissipate quite a bit of power. For example, if the motor windings have a resistance of 5 ohms and a rated current of 1 amp, and a 25 volt power supply is used, T1 plus R1 will dissapate, between them, 20 watts! The circuits discussed in the following sections avoid this waste of power while retaining the performance advantages of the circuit given here.

 

When an H-bridge bipolar drive is used with a resistive current limiter, as shown in Figure 4.1, the resistor R2 was not needed because current could flow backwards through R1. When a transistorized current limiter is used, current cannot flow backwards through T1, so a separate current path back to the positive supply must be provided to handle the decaying current through the motor windings when the switches are opened. R2 serves this purpose here, but a zener diode may be substituted to provide even faster turn-off.

 

The performance of a motor run with a current limited power supply is noticably better than the performance of the same motor run with a resistively limited supply, as illustrated in Figure 4.4:

 

 

 

 

 

 

Figure 4.4

 

With either a current limited supply or a resistive current limiter, the initial rate of increase of the current through the inductive motor winding when the power is turned on depends only on the inductance of the winding and the supply voltage. As the current increases, the voltage drop across a resistive current limiter will increase, dropping the voltage applied to the motor winding, and therefore, dropping the rate of increase of the current through the winding. As a result, the current will only approach the rated current of the motor winding asymptotically In contrast, with a pure current limiter, the current through the motor winding will increase almost linearly until the current limiter cuts in, allowing the current to reach the limit value quite quickly. In fact, the current rise is not linear; rather, the current rises asymptotically towards a limit established by the resistanc
e of the motor winding and the resistance of the sense resistor in the current limiter. This maximum is usually well above the rated current for the motor winding.

 

 

Open Loop Current Limiters

 

Both the resistive and the linear transistorized current limiters discussed above automatically limit the current through the motor winding, but at a considerable cost, in terms of wasted heat. There are two schemes that eliminate this expense, although at some risk because of the lack of feeback about the current through the motor.

 

 

Use of a Voltage Boost

 

If you plot the voltage across the motor winding as a function of time, assuming the use of a transistorized current limiter such as is illustrated in Figure 4.3, and assuming a 1 amp 5 ohm motor winding, the result will be something like that illustrated in Figure 4.5:

 

 

 

 

 

 

 

 

 

 

 

Figure 4.5

 

As long as the current is below the current limiter’s set point, almost the full supply voltage is applied across the motor winding. Once the current reaches the set point, the voltage across the motor winding falls to that needed to sustain the current at the set point, and when the switches open, the voltage reverses briefly as current flows through the diode network and R2. An alternative way to get this voltage profile is to use a dual-voltage power supply, turning on the high voltage for as long as it takes to bring the current in the motor winding up to the rated current, and then turning off the high voltage and turning on the sustaining voltage. Some motor controllers do this directly, without monitoring the current through the motor windings. This provides excellent performance and minimizes power losses in the regulator, but it offers a dangerous temptation.

 

If the motor does not deliver enough torque, it is tempting to simply lengthen the high-voltage pulse at the time the motor winding is turned on. This will usually provide more torque, although saturation of the magnetic circuits frequently leads to less torque than might be expected, but the cost is high! The risk of burning out the motor is quite real, as is the risk of demagnitizing the motor rotor if it is turned against the imposed field while running hot. Therefore, if a dual-voltage supply is used, the temptation to raise the torque in this way should be avoided!

 

The problems with dual voltage supplies are particularly serious when the time intervals are under software control, because in this case, it is common for the software to be written by a programmer who is insufficiently aware of the physical and electrical characteristics of the control system.

 

 

Use of Pulse Width Modulation

 

Another alternative approach to controlling the current through the motor winding is to use a simple power supply controlled by pulse width modulaton (PWM) or by a chopper. During the time the current through the motor winding is increasing, the control system leaves the supply attached with a 100% duty cycle. Once the current is up to the full rated current, the control system changes the duty cycle to that required to maintain the current. Figure 4.6 illustrates this scheme:

 

 

 

 

 

 

Figure 4.6

For any chopper or pulse width modulator, we can define the duty-cycle D as the fraction of each cycle that the switch is closed:

D = Ton / (Ton + Toff)

Where

Ton — time the switch is closed during each cycle

Toff — time the switch is open during each cycle

The voltage curve shown above indicates the full supply voltage being applied to the motor winding during the on-phase of every chopper cycle, while when the chopper is off, a negative voltage is shown. This is the result of the forward voltage drop in the diodes that are used to shunt the current when the switches turn off, plus the external resistance used to speed the decay of the current through the motor winding. For large values of Ton or Toff, the exponential nature of the rise and fall of the current through the motor winding is significant, but for sufficiently small values, we can approximate these as linear. Assuming that the chopper is working to maintain a current of I and that the amplitude is small, we will approximate the rates of rise and fall in the current in terms of the voltage across the motor winding when the switch is closed and when it is open:

 

Von = Vsupply – I(Rwinding + Ron)

Voff = Vdiode + I(Rwinding + Roff)

Here, we lump together all resistances in series with the winding and power supply in the on state as Ron, and we lump together all resistances in the current recirculation path when the switch(es) are open as Roff. The forward voltage drops of any diodes in the current recirculation path have been lumped as Vdiode; if the off-state recirculation path runs from ground to the power supply (H-bridge fast decay mode), the supply voltage must also be included in Vdiode. Forward voltage drops of any switches in the on-state and off-state paths should also be incorporated into these voltages.

To solve for the duty cycle, we first note that:

 

dI/dt = V/L

Where

I — current through the motor winding

V — voltage across the winding

L — inductance of the winding

 

We then substitute the specific voltages for each phase of operation:

 

Iripple / Toff = Voff / L

Iripple / Ton = Von / L

Where

Iripple — the peak to peak ripple in the current

Solving for Toff and Ton and then substituting these into the definition of the duty cycle of the chopper, we get:

D = Ton / (Ton + Toff) = Voff / (Von + Voff)

If the forward voltage drops in diodes and switches are negligable, and if the only significant resistance is that of the motor winding itself, this simplifies to:

D = I Rwinding / Vsupply = Vrunning / Vsupply

This special case is particularly desirable because it delivers all of the power to the motor winding, with no losses in the regulation system, without regard for the difference between the supply voltage and the running voltage.

The AC ripple Iripple superimposed on the running current by a chopper can be a source of problems; at high frequencies, it can be a source of RF emissions, and at audio frequencies, it can be a source of annoying noise. For example, with audio frequency chopping, most stepper controlled systems will “squeel”, sometimes loudly, when the rotor is displaced from the equilibrium position. For small systems, this is usually no more than a minor nuisance, but in systems with large numbers of high power steppers, the ripple currents can induce dangerous AC voltages on nearby signal lines and dangerous currents in nearby ground lines. To find the ripple amplitude, first recall that:

 

Iripple / Toff = Voff / L

Then solve for Iripple:

Iripple = Toff Voff / L

Thus, to reduce the ripple amplitude at any particular duty cycle, it is necessary to increase the chopper frequency. This cannot be done without limit because switching losses increase with frequency. Note that this change has no significant effect on AC losses; the decrease in such losses due to decreased amplitude in the ripple is generally offset by the effect of increasing frequency.

The primary problem with use of a simple chopping or pulse-width modulation control scheme is that it is completely open loop. Design of good chopper based control systems requires knowledge of motor characteristics such as inductance that are frequently poorly documented, and as with dual-voltage supplies, when motor performance is marginal,
it is very tempting to increase the duty-cycle without attention to the long-term effects of this on the motor. In the designs that follow, this weakness will be addressed by introducing feedback loops into the low level drive system to directly monitor the current and determine the duty cycle.

 

One-Shot Feedback Current Limiting

 

The most common approach to automatically adjusting the duty cycle of the switches in the stepper driver involves monitoring the current to the motor windings; when it rises too high, the winding is turned off for a fixed interval. This requires a current sensing system and a one-shot, as illustrated in Figure 4.7:

 

 

 

 

 

Figure 4.7

Figure 4.7 illustrates a unipolar drive system. As with the circuit given in Figure 4.3, R1 should be as small as possible, limited only by the requirement that the sense voltage provided to the comparator must be high enough to be within its operating range. Note that when the one-shot output (¬Q) is low, the voltage across R1 no-longer reflects the current through the motor winding. Therefore, the one-shot must be insensitive to the output of the comparator between the time it fires and the time it resets. Practical circuit designs using this approach involve some complexity to meet this constraint! Selecting the value of R2 for the circuit shown in Figure 4.7 poses problems. If R2 is large, the current through the motor windings will decay quickly when the higher level control system turns off this motor winding, but when the winding is turned on, the current ripple will be large and the power lost in R2 will be significant. If R2 is small, this circuit will be very energy efficient but the current through the motor winding will decay only slowly when this winding is turned off, and this will reduce the cutoff speed for the motor.

 

The peak power dissipated in R2 will be I2R2 during Toff and zero during Ton; thus, the average power dissipated in R2 when the motor winding is on will be:

 

P2 = I2R Toff / (Ton + Toff)

Recall that the duty cycle D is defined as Ton/(Ton+Toff) and may be approximated as Vrunning/Vsupply. As a result, we can approximate the power dissipation as:

P2 = I2R2 (1 – Vrunning/Vsupply).

Given the usual safety margins used in selecting power resistor wattages, a better approximation is not necessary.

When designing a control system based on pulse width modulation, note that the cutoff time for the one-shot determines Toff, and that this is fixed, determined by the timing network attached to the one-shot. Ideally, this should be set as follows:

 

Toff = L Iripple / Voff

This presumes that the inductance L of the motor winding is known, that the acceptable magnitude of Iripple is known, and that Voff, the total reverse voltage in the current recirculation path, is known and fixed. Note that this scheme leads to a variable chopping rate. As with the linear current limiters shown in Figure 4.3, the full supply voltage will be applied during the turn-on phase, and the chopping action only begins when the motor winding reaches the current limit set by Vref. This circuit will vary the chopping rate to compensate for changes in the back EMF of the motor winding, for example, those caused by rotor motion; in this regard, it offers the same quality of regulation as the linear current limiter. The one-shot current regulator shown in Figure 4.7 can also be applied to an H-bridge regulator. The encoded H-bridge shown in Figure 3.13 is an excellent candidate for this application, as shown in Figure 4.8:

 

Figure 4.8

Unlike the circuit in Figure 4.7, this circuit does not provide design tradeoffs in the selection of the resistance in the current decay path; instead, it offers the same selection of decay paths as was available in the original circuit from Figure 3.13. If the X and Y control inputs are held in a running mode (01 or 10), the current limiter will alternate between that running and slow decay modes, maximizing energy efficiency. When the time comes to turn off the current through the motor winding, the X and Y inputs may be set to 00, using fast decay mode to maximize the cutoff speed, while if the damping effect of dynamic braking is needed to control resonance, X and Y may be set to 11.

Note that the current recirculation path during dynamic braking does not pass through R1, and as a result, if the motor generates a large amount of power, burnt out components in the motor or controller are likely. This is unlikely to cause problems with stepping motors, but when dynamic braking is used with DC motors, the current limiter should be arranged to remain engaged while in braking mode!

 

 

Practical Examples

SGS-Thompson (and others) L293 (1A) and L298 (2A) dual H-bridges are designed for easy use with partial feedback current limiters. These chips have enable inputs for each H-bridge that can be directly connected to the output of the one-shot, and they have ground connections for motor-power that are isolated from their logic ground connections; this allows sense resistors to be easily incorporated into the circuit. The 3952 H-bridge from Allegro Microsystems can handle up to 2-amps at 50 volts and incorporates all of the logic necessary for current control, including comparators and one-shot. This chip is available in many package styles; Figure 4.9 illustrates the DIP configuration wired for a constant current limit:

 

 

 

 

 

 

Figure 4.9

If Rt is 20 Kohm, and Ct is 1000pF, Toff for the pulse-width modulation will be fixed at 20 (±2) microseconds. The 3952 chip incorporates a 10 to 1 voltage divider on the Vref input, so attaching Vref to the 5 volt logic supply sets the actual reference voltage to 0.5 V. Thus, if the sense resistor Rs is 0.5 ohms, this arrangement will attempt to maintain a regulated current through the load of 1 A.

Note that all power switching chips are potentially serious sources of electromagnetic interfence! The 47µF capacitor shown between the motor power and ground should be as close to the chip as possible, and the path from the SENSE pin through Rs to ground and back to a ground pin of the chip should be very short and with a very low resistance.

 

On the 5 volt side, because Vref is taken from Vcc, a small decoupling capacitor should be placed very close to the chip. It may even be appropriate to isolate the Vref input from Vcc with a small series resistor and a separate decoupling capacitor. If this is done, note that the resistance from the Vref pin to ground through the chip’s internal voltage divider is around 50 Kohms.

 

One of the more dismaying features of the 3952 chip, as well as many of its competitors, is the large number of control inputs. These are summarized in the following table:

 

BRAKE ENABLE PHASE  MODE  OUTa OUTb Notes 

       

0  -  -  0   0  0  Brake 

0  -  -  1   0  0  Limited Brake 

       

1  1  -  0   -  -  Standby 

1  1  -  1   -  -  Sleep 

       

1  0  0  0   0  1  Reverse, Slow 

1  0  0  1   0  1  Reverse, Fast 

1  0  1  0   1  0  Forward, Slow 

1  0  1  1   1  0  Forward, Fast 

 

In the forward and reverse running modes, the mode input determines whether fast or slow decay modes are used during Toff. In the dynamic braking modes, the mode input determines whether the current limiter is enabled. This is of limited value with stepping motors, but use of dynamic braking without a current limiter can be dangerous with DC motors. In sleep mode, the power consumption of the chip is minimized. From the perspect
ive of the load, sleep and standby modes put the load into fast decay mode (all switches off) but in sleep mode, the chip draws considerably less power, both from the logic supply and the motor supply.

 

 

Hysteresis Feedback Current Limiting

In many cases, motor control systems are expected to operate acceptably with a number of different stepping motors. The one-shot based current regulators illustrated in Figures 4.7 to 4.9 have an accuracy that depends on the inductance of the motor windings. Therefore, if fixed accuracy is required, any motor substation must be balanced by changes to the RC network that determines the off-time of the one-shot.

 

This section deals with alternative designs that eliminate the need for this tuning. These alternative designs offer fixed precision current regulation over a wide range of load inductances. The key to this approach is arrange the recirculation paths so that the current-sense resistor R1 is always in the circuit, and then turn the switches on or off depending only on the current.

 

The usually way to build this type of controller is to use a comparator with a degree of hysteresis, for example, by feeding the output of the comparator back into one of its inputs through a resistor network, as illustrated in Figure 4.10:

 

 

Figure 4.10

To compute the desired values of R2 and R3, we note that:

Vripple > Vhysteresis

Where:

Vripple = Iripple R1

Iripple — the maximum ripple allowed in the current

and:

Vhysteresis = Vswing R2 / (R2 + R3)

Vswing — the voltage swing at the output of the comparator

We can solve this for the ratio of the resistances:

R2 / (R2 + R3) < Iripple R1 / Vswing

For example, if R1 is 0.5 ohms and we wish to regulate the current to within 10 milliamps, using a comparator with TTL compatable outputs and a voltage swing of 4 volts, the ratio must be no greater than .00125.

Note that the sum R2 + R3 determines the loading on Vref, assuming that the input resistance of the comparator is effectively infinite. Typically, therefore, this sum is made quite large.

 

One problem with the circuit given in Figure 4.10 is that it does not limit the current through the motor in dynamic braking or slow decay modes. Even if the current through the sense resistor vastly exceeds the desired current, switches B and D will remain closed in dynamic braking mode, and if the reference voltage is variable, rapid drops in the reference voltage will not be enforced by this control system.

 

The designers of the Allegro 3952 chip faced this problem, and passed the solution back to the user, providing a MODE input to determine whether the chopper alternated between running and fast decay mode or running and slow decay mode. Note that this chip uses a fixed off-time set by a one-shot, and therefore, switching between the two decay modes will change the precision of the current regulator. Given that such a change in precision is acceptable, we can modify the circuit from Figure 4.10 to automatically thrown the system into fast-decay mode if the running or dynamic braking current exceeds the set-point of the comparator by too great a margin. Figure 4.11 illustrates how this can be done using a second comparator:

 

Figure 4.11

As shown in Figure 4.11, the lower comparator directly senses the voltage across R1, while the upper comparator senses a higher voltage, determined by a resistor network. This network should hold the negative inputs of the two comparators just far enough apart to guarantee that, as the voltage across R1 rises, the top comparator will always open the top switches before the bottom comparator opens the bottom switches, and as the voltage across R1 falls, the bottom comparator will always close the bottom switches before the top comparator closes the top switches.

As a result, this system has two basic steady-state running modes. If the motor winding is drawing power, one of the bottom switches will remain closed while the opposite switch on the top is used to chop the power to the motor winding, alternating the state of the system between running and slow-decay mode.

 

If the motor winding is generating power, the top switches will remain open and the bottom switches will do the chopping, alternating between fast-decay and slow-decay modes as needed to keep the current within limits. If the two comparators have accuracies on the order of a millivolt with hysteresis on the order of 5 millivolts, it is reasonable to use a 5 millivolt difference between the top and bottom comparator. If we use the 5 volt logic supply as the pull-up supply for the resistor network, and we assume a nominal operating threshold of around 0.5 volts, the resistor network should have a ratio of 1:900; for example, a 90k resistor from +5 and a 100 ohm resistor between the two comparator inputs.

 

 

Practical Examples

 

The basic idea described in this section is also applicable to unipolar stepping motor controllers, although in this context, it is somewhat easier to apply if the reference voltage is measured with respect to the unregulated motor power supply. Figure 4.12 illustrates a practical example, using the forward voltage drop across an ordinary silicon diode as the reference voltage.

 

 

Figure 4.12

The circuit shown in Figure 4.12 uses a 2.4K resistor to provide a bias current of 10ma to the reference diode. A small capacitor should be added across the reference diode if the motor power supply is minimally regulated.

 

The 0.6 ohm value used for the current sensing resistor sets the regulator to 1 amp, assuming that the reference voltage is 0.6 volts. The 1000 to 1 ratio on the feedback network around the comparator sets the allowed ripple in the regulated current to around 8 ma.

               

The comparator shown in Figure 4.12 can be powered from the minimally regulated motor power supply, but only if it is able to operate with the inputs very close to its positive supply voltage. Although I have not tried it, the Mitsubishi M5249L comparator appears to be ideally suited to this job; it can work from a positive supply of up to 40 volts, and the input voltages are allowed to slightly exceed the positive supply voltage! The output of this comparator is open collector, so the hysteresis network shown in the figure also acts as a pull-up network, providing a pull-up current of a few milliamps. The diode to +5 shown in the figure clamps the comparator output to the logic supply voltage, protecting the and gate inputs from overvoltage.

 

 

Other Current Sensing Technologies

 

The feedback loops of all of the current limiters given above use the voltage drop across a small resistor to measure the current. This is an excellent choice for small motors, but it poses difficulties for large high-current motors! There are other current sensing technologies appropriate for such settings, most notably those that deliver only a fraction of the motor current to the sensing resistor, and those that measure the current by sensing the magnetic field around the conductor.

 

National Semiconductor had incorporated a very clever current sensor into a number of their H-bridges. This delivers a current to the sense resistor that is proportional to the current through the motor winding, but far lower. For example, on the LMD18200 H-bridge, the sense resistor receives exactly 377 micro amps per ampere flowing through the motor winding.

 

The key to the current sensing technology used in the National Semiconductor line of H-bridges is found in the internal structure of the DMOS power switching transistors they use. These transistors are composed of thousan
ds of small MOSFET transistor cells wired in parallel. A small but representative fraction of these cells, typically 1 in 4000, is used to extract the sense current while the remainder of the cells controls the motor current. The data sheet for the National LMD18245 H-bridge contains an excellent write-up on how this is done.

 

When very high currents are involved, precluding use of an integrated H-bridge, an appealing and well established current sensing technology involves the use of a split ferrite core and a hall effect sensor, as illustrated in Figure 4.13:

 

Figure 4.13

Simple linear Hall effect sensors require a small regulated bias current between two of their terminals, and they generate a DC voltage proportional to the magnetic field on a third terminal. The magnetic field across the gap sawed in the ferrite core is proportional to the current through the wire, and therefore, the voltage reported by the Hall effect sensor will be proportional to the current.

Allegro Microsystems and others make a full lines of Hall effect sensors, but pre-calibrated hall effect current sensors are available; these include the split core, the hall effect sensor, and auxiliary components, all mounted on a small PC board or potted as a unit. Newark Electronics lists a few sources of these, including Honeywell, F. W. Bell and LEM Instruments.

 

An intriguing new current sensor is just becoming available, as of 1998, based on a thin-film magneto resistive sensor; the sensitivity of this technology eliminates the need for the ferrite core and the result is a very compact current sensor. The NT series sensors made by F. W. Bell use this technology.

 

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1. Introduction

The principle of identifying critical success factors as a basis for determining the information needs of managers was proposed by RH Daniel (1961 Harvard Business Review – HBR) as an interdisciplinary approach with a potential usefulness in the practice of evaluation within library and information units but popularized by F Rockart (1979 Harvard Business Review – HBR

The following as an example of generic CSF’s:

New product development, Good distribution, and Effective advertising

2. Five key sources of Critical Success Factors

MAIN ASPECTS OF Critical Success Factors and their use in analysis

CSF’s are tailored to a firm’s or manager’s particular situation as different situations (e.g. industry, division, individual) lead to different critical success factors. Rockart and Bullen presented five key sources of CSF’s:

The industry, Competitive strategy and industry position, Environmental factors, Temporal factors, and Managerial position (if considered from an individual’s point of view). Each of these factors is explained in greater detail below.

3. Basic Type of CSFs

There are four basic types of CSFs according to Rockart. They are:

Industry CSFs resulting from specific industry characteristics; Strategy CSFs resulting from the chosen competitive strategy of the business; Environmental CSFs resulting from economic or technological changes; and Temporal CSFs resulting from internal organizational needs and changes.

4. Critical Success Factor Method



Start with a vision Mission statement Develop 5-6 high level goals Develop hierarchy of goals and their success factors Lists of requirements, problems, and assumptions Leads to concrete requirements at the lowest level of decomposition (a single, implementable idea) Along the way, identify the problems being solved and the assumptions being made Cross-reference usage scenarios and problems with requirements Analysis matrices Problems vs. Requirements matrix Usage scenarios vs. Requirements matrix Solid usage scenarios Relationship to Usage Scenarios Usage scenarios or “use cases”; provide a means of determining: Are the requirements aligned and self-consistent? Are the needs of the user being met as well as those of the enterprise? Are the requirements complete Results of the Analysis

5. Example of Critical Success factors for Company XYZ

Critical Success Factor

Source of

CSF

Primary Measures

& Targets

1. Increase Number of customers

Industry

95% customer retention rate;

15% new customers per yr

2. Instal PC-based customer service

…hot line

Strategy

90% of customer queries

answered in 1 hour

3. Increase number of customer service reps

Strategy

3 reps per 100 customers

4. Restructure capital structure

Environmental

Lower cost of capital by 2%

5. Raise employee morale and

…productivity

Temporal

Increase employee retention

rate to 95% / yr.

References

Porter, Michael E., ed., Competition in Global Industries, Boston, Mass.: Harvard Business School Press, 1986. Porter, Michael E., Competitive Advantage: Creating and Sustaining Superior Performance, New York: The Free Press, 1985. Porter, Michael E., Competitive Advantage of Nations, New York: The Free Press, 1990

John S. Reel, “Critical Success Factors In Software Projects,” IEEE Software, vol. 16, no. 3, pp. 18-23, May/June 1999, doi:10.1109/52.765782

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With easy to use Internet access software, you have access to lectures, questions and assignments from your professors and then you can print them out and review them off-line. You will also have access to a full range of online research libraries and services.

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